I. Field of the Invention
The present invention relates to communications. More particularly, the present invention relates to a novel and improved quadrature modulator and demodulator.
II. Description of the Related Art
In many modern communication systems, digital transmission is utilized because of improved efficiency and the ability to detect and correct transmission errors. Exemplary digital transmission formats include binary phase shift keying (BPSK), quaternary phase shift keying (QPSK), offset quaternary phase shift keying (OQPSK), m-ary phase shift keying (m-PSK), and quadrature amplitude modulation (QAM). Exemplary communication systems which utilize digital transmission include code division multiple access (CDMA) communication systems and high definition television (HDTV) systems. The use of CDMA techniques in a multiple access communication system is disclosed in U.S. Pat. No. 4,901,307, entitled "SPREAD SPECTRUM MULTIPLE ACCESS COMMUNICATION SYSTEM USING SATELLITE OR TERRESTRIAL REPEATERS", and U.S. Pat. No. 5,103,459, entitled "SYSTEM AND METHOD FOR GENERATING WAVEFORMS IN A CDMA CELLULAR TELEPHONE SYSTEM", both assigned to the assignee of the present invention and incorporated by reference herein. An exemplary HDTV system is disclosed in U.S. Pat. No. 5,452,104, U.S. Pat. No. 5,107,345, and U.S. Pat. No. 5,021,891, all three entitled "ADAPTIVE BLOCK SIZE IMAGE COMPRESSION METHOD AND SYSTEM", and U.S. Pat. No. 5,576,767, entitled "INTERFRAME VIDEO ENCODING AND DECODING SYSTEM", all four patents are assigned to the assignee of the present invention and incorporated by reference herein.
In the CDMA system, a base station communicates with one or more remote stations. The base station is typically located at a fixed location. Thus, power consumption is less important consideration in the design of the base station. The remote stations are typically consumer units which are produced in high quantity. Thus, cost and reliability are important design considerations because of the number of units produced. Furthermore, in some applications such as a CDMA mobile communication system, power consumption is critical because of the portable nature of the remote station. Tradeoffs between performance, cost, and power consumption are usually made in the design of the remote stations.
In digital transmission, the digitized data is used to modulated a carrier sinusoid using one of the formats listed above. The modulated waveform is further processed (e.g. filtered, amplified, and upconverted) and transmitted to the destination device. At the destination device, the transmitted RF signal is received and demodulated by a receiver.
A block diagram of an exemplary transmitter 100 of the prior art which is used for quadrature modulation of QPSK, OQPSK, and QAM signals is illustrated in FIG. 1A. Transmitter 100 can be used at the base station or the remote station. Within quadrature modulator 110a of transmitter 100, the inphase (I) and quadrature (Q) signals are provided to mixers 112a and 112b which modulate the signals with the inphase and quadrature intermediate frequency (IF) sinusoids, respectively. Quadrature splitter 114 receives the IF sinusoid (IF LO) and provides the inphase and quadrature IF sinusoids which are approximately equal in amplitude and 90 degrees out of phase with respect to each other. The modulated I and Q signals from mixers 112a and 112b are provided to summer 116 and combined. In many applications, the signal from summer 116 is provided to mixer 118 which upconverts the signal to the desired frequency with the radio frequency (RF) sinusoid (RF LO). Although not shown in FIG. 1A for simplicity, filtering and/or amplification can be interposed between successive stages of summers and mixers.
The modulated signal from mixer 118 is provided to filter 130 which filters out undesirable images and spurious signals. The filtered signal is provided to amplifier (AMP) 132 which amplifies the signal to produce the required signal amplitude. The amplified signal is routed through duplexer 134 and transmitted from antenna 136 to the destination device.
A block diagram of an exemplary direct quadrature modulator 110b is shown in FIG. 1B. Within direct quadrature modulator 110b, the I and Q signals are provided to mixers 152a and 152b which modulate the signals with the inphase and quadrature RF sinusoids, respectively. Quadrature splitter 154 receives the direct RF sinusoid (direct RF LO) and provides the inphase (I LO) and quadrature (Q LO) sinusoids which are approximately equal in amplitude and 90 degrees out of phase with respect to each other. The modulated I and Q signals from mixers 152a and 152b are provided to summer 156 and combined to provide the modulated signal.
Quadrature modulator 110a performs modulation using a two steps process whereby quadrature modulation is performed at an IF frequency and upconverted to the desired RF frequency. Quadrature modulator 110a offers several advantages. First, quadrature splitter 114 can be more easily designed and manufactured to meet the required specification at the lower IF frequency. Second, the two sinusoids design (IF LO and RF LO) offers flexibility in the frequency plan and simplification of the filtering.
Direct quadrature modulator 110b performs the same functions as quadrature modulator 110a. However, direct quadrature modulator 110b performs modulation directly at the desired RF frequency using a single step process, thereby eliminating the upconversion step. The simplicity in the design of modulator 110b is offset by the performance requirements of quadrature splitter 154. In particular, it is much more difficult to design and manufacture quadrature splitter 154 having the required amplitude balance and quadrature phase at the higher RF frequency.
A method for generating inphase and quadrature sinusoids at RF frequency having the required performance is disclosed in U.S. Pat. No. 5,412,351, entitled "QUADRATURE LOCAL OSCILLATOR NETWORK", and incorporated by reference herein. A block diagram of quadrature local oscillator network 170 as disclosed in U.S. Pat. No. 5,412,351 is shown in FIG. 1C. Within quadrature local oscillator network 170, the IF sinusoid is provided to quadrature splitter 172 which provides the inphase and quadrature IF sinusoids. The inphase IF sinusoid is provided to mixers 176a and 176d and the quadrature IF sinusoid is provided to mixers 176b and 176c. Similarly, the RF sinusoid is provided to quadrature splitter 174 which provides the inphase and quadrature RF sinusoids. The inphase RF sinusoid is provided to mixers 176b and 176d and the quadrature RF sinusoid is provided to mixers 176a and 176c. Mixers 176a and 176b mix the two input signals and provide the upconverted signals to summer 178a which combines the signals to provide the inphase direct sinusoid (I LO). Similarly, mixers 176c and 176d mix the two input signals and provide the upconverted signals to summer 178b which combines the signals to provide the quadrature direct sinusoid (Q LO). The inphase and quadrature direct sinusoids can be provided to mixers 152a and 152b, respectively, as shown in FIG. 1B.
Ideally, the inphase and quadrature sinusoids from a phase splitter are equal in amplitude and 90 degrees out of phase with respect to each other. At the RF frequency, this is difficult to achieve. For ideal quadrature splitters 172 and 174 (with no amplitude imbalance and no phase error), the inphase (I LO) and quadrature (Q LO) sinusoids are exactly equal in amplitude and 90 degree out of phase with respect to each other. Each sinusoid comprises a single tone at the difference frequency (f.sub.RF -f.sub.IF) and no other mixing terms. The I LO and Q LO can be expressed as: EQU I.sub.-- LO(t)=cos(.omega..sub.RF -.omega..sub.IF)t EQU Q.sub.-- LO(t)=sin(.omega..sub.RF -.omega..sub.IF)t (1)
Although quadrature local oscillator network 170 in FIG. 1C is configured to produce sinusoids at the difference frequency (f.sub.RF -f.sub.IF), network 170 can also be reconfigured to produce sinusoids at the sum frequency (f.sub.RF +f.sub.IF).
Quadrature local oscillator network 170 generates inphase and quadrature sinusoids which have improved performance over sinusoids generated by other quadrature splitters of the prior art. In particular, quadrature local oscillator network 170 substantially reduces the sensitivity of the output sinusoids to amplitude imbalance and/or phase error in quadrature splitters 172 and 174. Amplitude imbalance and/or phase error in quadrature splitters 172 and 174 do not substantially affect the amplitude balance and quadrature phase of the output sinusoids. Instead, amplitude imbalance and phase error of quadrature splitters 172 and 174 manifest themselves as spurious signals which can be filtered. For example, an amplitude imbalance of .DELTA. at an output of quadrature splitter 172 or 174 results in I LO and Q LO sinusoids which can be expressed as: ##EQU1##
As used in this specification, an amplitude imbalance of .DELTA. denotes that one output sinusoid from a quadrature splitter has an amplitude of 1 and the other output sinusoid has an amplitude of (1+.DELTA.). From equation (2), each output from network 170 comprises the desired sinusoid and a spurious signal. The spurious signal has an amplitude of half the amplitude error (.DELTA./2) and is located at 2f.sub.IF from the desired sinusoid. This spurious signal is small in amplitude and can be filtered. More importantly, notice that the desired output sinusoids from network 170 are still amplitude balanced and in quadrature phase with each other.
A phase error of .phi. at an output of quadrature splitter 172 or 174 results in I LO and Q LO sinusoids which can be expressed as: ##EQU2##
As used in this specification, a phase error .phi. denotes that the phase of the quadrature sinusoid is (90.sup.0.+-..phi.) with respect to the phase of the inphase sinusoid. From equation (3), notice that the phase error .phi. results in each output from network 170 comprising the desired sinusoid and two spurious signals having amplitudes of [1/2-cos(.phi.)/2] and [sin(.phi.)/2] and located at 2f.sub.IF from the desired sinusoid. For small phase error .phi., the spurious signals are small in amplitude. In addition, the spurious signals can be filtered since they are located at 2f.sub.IF from the desired sinusoid. Each output from network 170 also comprises a small quadrature component of the desired sinusoid having an amplitude of sin (.phi.)/2. This quadrature component causes a slight rotation in the phase of the output sinusoid. However, since the inphase and quadrature output sinusoids comprise quadrature components having the sample amplitude {sin(.phi.)/2}, the 90 degree phase difference between the output sinusoids is maintained.
Although quadrature local oscillator network 170 provides the requisite performance, a major disadvantage is the power consumption. Notice in FIG. 1C that all four mixers 176 and both summers 178 operate at the RF frequency. To achieve the required circuit performance (e.g., bandwidth and linearity) at RF frequency, these circuits are biased with high current. For some applications, such as CDMA communication system, power consumption is a critical design parameter. There exists a long felt need in the industry to provide a quadrature modulator and demodulator which provide the requisite level of performance while minimizing power consumption.